User GuideSLUU087B - September 200110-Watt Flyback Converter Using the UCC3809Power Supply Control ProductsContents123456710111213141516171819Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2Description. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5Theory of Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5UCC3809EVM–052 Operating Instructions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6Determine the Maximum on Time (DMAX) for the Primary Switch. . . . . . . . . . . . . . . . . . . . . . . . 7Determine the Turns Ratio of the Flyback Inductor. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7Primary Inductance. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7Soft Start. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8Self-Biasing Winding. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8Selection of the Switching Element. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8Current Sense. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9Gate Drive. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10Output Diode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10Output and Input Capacitors. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11Loop Compensation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11Slope Compensation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13Silkscreen. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13Conclusion. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 131SLUU087B11.1Introduction Design ProcedureThe UCC3809EVM evaluation module is an isolated 10-W, 3.3-V converter designed for telecom input voltages,nominally –48 Vdc but ranging from –32 Vdc to –75 Vdc. The UCC3809 economy primary-side controller is usedas the pulse-width modulator (PWM), combined with a TLV431 low-voltage adjustable-precision shunt regulatoron the secondary side. The nominal switching frequency for this converter is approximately 400 kHz. TheUCC3809 is configured with a 50% duty-cycle clamp.1.2Operating SpecificationsSPECIFICATIONInput voltage rangeOutput voltage rangeOutput current rangeOperating frequencyOutput rippleEfficiency (VIN = 48 V, IOUT = 3 A)MIN323.2180344TYP483.338075MAX753.3833.3420100UNITSVVAkHzmV%The circuit, shown in Figure 1, is a discontinuous fixed-frequency current-mode controlled flyback converter.This converter contains features such as external shut down, optocoupler isolation, programmable maximumduty cycle, and a summing node for voltage feedback, current sense, and slope compensation.Table 1. UCC3809 10-W Flyback Converter Bill of MaterialsReferenceC1C2C3C4C5C6C7CapacitorC8C9C10, C11, C12C13C14C15C16, C17C18C19QTY1111111113111211DescriptionCeramic, 39 pF, ±5%, 50 V, NPO, 0603Ceramic, 1.0 µF, +80%/–20%, 50 V, Y5V, 0805Ceramic, 10 nF, ±10%, 50 V, X7R, 0603Ceramic, 10 µF, ±10%, 35 V, Y5V, 1210Ceramic, 4.7 nF, ±10%, 50 V, X7R, 0603Ceramic, 120 pF, ±5%, 50 V, NPO, 0603Ceramic, 1.5 nF, ±10%, 50 V, X7R, 0603Ceramic, 150 pF, ±5%, 50 V, NPO, 0603Tantalum, 470 µF, ±10%, 6.3 V, 7343HCeramic, 47 pF, ±5%, 50 V, NPO, 0603Ceramic, 10 µF, ±10%, 6.3 V, X5R, 1206Ceramic, 0.1 µF, +80%/–20%, 16V, Y5V, 0603Ceramic, 220 pF, ±5%, 50 V, NPO, 0603Ceramic, 1 µF, ±10%, 100 V, X7R, 2225Ceramic, 1 µF, ±10%, 16 V, X7R, 1206ManufacturerPanasonicTaiyo YudenPanasonicTaiyo YudenPanasonicPanasonicPanasonicPanasonicKemetPanasonicTaiyo YudenPanasonicPanasonicKemetPanasonicPart NumberECJ–1VC1H390JUMK212F105ZGECU–V1H103KBVECJ–1VF1C474ZGMK325F106ZHECU–V1H472KBVECJ–1VC1H121JECU–V1H152KBVECU–V1H151JCVT510X477K006AS *ECJ–1VC1H470JJMK316BJ106MLECJ–1VF1C104ZECU–V1H221JCVC2225C105K1RAC *ECJ–3YB1C105KCeramic, 0.47 µF, +80%/–20%, 16 V, Y5V, 0603PanasonicNOTE:* Equivalent substitution not recommended210-Watt Flyback Converter Using the UCC3809SLUU087BReferenceD1QTY1DescriptionFast Switching, 70 V, 10 mA, SOT–23ManufacturerVishay/LiteOn PowerSemiconductor(Diodes Inc.)Vishay/LiteOn PowerSemiconductor(Diodes Inc.)International RectifierDiodes, Inc.MotorolaVishay/LiteOn PowerSemiconductor(Diodes Inc.)International RectifierPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPanasonicPulseTexas InstrumentsTexas InstrumentsMill–MaxPart NumberBAV70DIDiodeD2D3D4111121212111111111111111111111151Ultra-Fast, 200 V, 1 A, SMA/DO–214ACSCHOTTKY, 30 V, 12 A, TO–252AASCHOTTKY, 100 V, 1 A, SMA/DO–214AC846–01, STYLE 1NPN, 40 V, 350 mW, SOT–23N–Channel, 200 V, 4.8 A, 0.8 Ω, TO–252AAThick Film, 10.0 kΩ, ±1%, 1/16 W, 0603Thick Film, 5.11 kΩ, ±1%, 1/16 W, 0603Thick Film, 1.1 kΩ, ±1%, 1/16 W, 0603Thick Film, 10.0 Ω, ±1%, 1/16 W, 0603Thick Film, 0.39 Ω, ±5%, 1/2 W, 2010Thick Film, 82.5 kΩ, ±1%, 1/16 W, 0603Thick Film, 100 Ω, ±1%, 1/16 W, 0603Thick Film, 20 kΩ, ±5%, 1 W, 2512Thick Film, 150 Ω, ±1%, 1/4 W, 1210Thick Film, 5.1 Ω, ±5%, 1/4 W, 1210Thick Film, 15 kΩ, ±1%, 1/16 W, 0603Thick Film, 47.5 Ω,±1%, 1/16 W, 0603Thick Film, 619 kΩ, ±1%, 1/16 W, 0603Thick Film, 4.12 kΩ, ±1%, 1/16 W, 0603Thick Film, 6.81 kΩ, ±1%, 1/16 W, 0603Thick Film, 3.3 Ω, ±5%, 1/16 W, 0603Thick Film, 39.2 kΩ, ±1%, 1/16 W, 0603Thick Film, 392 Ω, ±1%, 1/16 W, 0603Thick Film, 49.9 Ω, ±1%, 1/16 W, 0603Thick Film, 3.92 kΩ, ±1%, 1/16 W, 0603Thick Film, 0 Ω, ±5%, 1/16 W, 0603Thick Film, 22.1 Ω, ±1%, 1/16 W, 060315 µH, RM5, 3F3Economy primary side controller, MSOP–8Low voltage adjustable precision shunt regulator, 1.25 V, SOT–23–5Terminal pinPCB, FR4, 4-Layer, 1 oz, 2.75””(L) x 1.00””(W)x 0.063””(T)”ES1DT12CWQ03FNB1100–13MOC207MMBT2222AIRFR220ERJ–3EKF1002VERJ–3EKF5111VERJ–3EKF1101VERJ–3EKF10R0VERJ–12ZQJR39UERJ–3EKF8252VERJ–3EKF1000VERJ–1WYJ203UERJ–14NF1500UERJ–14YJ5R1UERJ–3EKF1502VERJ–3EKF47R5VERJ–3EKF6193VERJ–3EKF4121VERJ–3EKF6811VERJ–3GSYJ3R3VERJ–3EKF3922VERJ–3EKF3920VERJ–3EKF49R9VERJ–3EKF3921VERJ–3GSY0R00VERJ–3EKF22R1VPB0028UCC3809P–2*TLV431ACDBVR*3156–2–00–01–00–00–08–0SLUP052OptoisolatorTransistorMOSFETQ1Q2, Q4Q3R1, R3R2R4, R6R5R7R8R9R10R11R12R13R14R15R16R17R18R19R20R21R22R23R24ResistorTransformerIntegratedCircuitTerminal pinT1U1Z1J1, J2, J3, J4, J5–NOTE:* Equivalent substitution not recommended 10-Watt Flyback Converter Using the UCC380934NOTE:NOTE:10-Watt Flyback Converter Using the UCC3809Figure 1. Flyback Converter Utilizing the UCC3809 Economy Primary-Side Controller.High-Temperature component. See EVM Warning and Restrictions at the back of this document.High-Voltage component. See EVM Warning and Restrictions at the back of this document.HIGH TEMPERATURE –SEE EVM WARNINGSR12 5.1AND RESTRICTIONSΩC81/4 W1.5 nFC10, C11, C12KemetD3470µF 6.3 VJ1T112CWQ03FN+PB00281,32T510X477(K)006ASJ3R11+R10C653,4C14VC18150 W20 kΩ4.7 nFIN = 32 – 75 V1µF1/4 W1 W610µFVOUT = 3.3 VC2225C105K1RAC–C711,2–120 pFD2R24D4J4J2ES1D22.1Ω23B1100–213C19.81µFR18D1C13C5 10µFHIGH VOLTAGE –R83.3ΩBAV70DICT47 pF25 V82.5 k7SEE EVM WARNINGSΩR5Q3AND RESTRICTIONS10ΩIRFR220D–Pak TO–252AAR61.1 kΩHIGH TEMPERATURE –R70.39ΩSEE EVM WARNINGSC11/2 WAND RESTRICTIONSR14R2139 pF47.9.9J5R96.3 V ΩΩSHUTDOWN100ΩUCC3809C175Q14R21FBREF8220 pF635.11 kΩR1939.2 kΩ2SSVDD772R17Q26.81 kΩMMBT2222AR223.92 kΩ3RT1OUT6R13R23811%R3C210 kΩ1.0µF15 kΩ0ΩMOC2071%4RT2GND525 VC15C4R1C160.1µF0.47µF10 kΩ6.3 VZ1R15R4619 kΩ220 pF1.1 kΩQ41%3TLV431MMBT2222A4C9150 pF 5%5REFR16C3 10 nF6.3 V4.12 kΩ6.3 VR201%392ΩUDG-00153SLUU087BSLUU087B2DescriptionA brief description of the circuit elements follows:DTransformer (a.k.a. flyback inductor) T1, transistor Q3, Schottky diode D3, input capacitor C18, and outputcapacitors C10 through C12 form the power stage of the converter. Power resistor R7 senses theprimary-switch current and converts this current into a voltage to be sensed by the primary-side controllerfeed-back comparator.DCapacitor C14 filters out high-frequency noise on the output bus directly at the output diode.DResistor R11 and capacitor C7, along with resistor R10, capacitor C6, and diode D2 make up the voltagesnubber and clamp, respectively, for the primary side; likewise, resistor R12 and capacitor C8 providesecondary-side snubbing.provided through the self-biasing (sometimes referred to as bootstrap) components D4 and C5.DResistor R8 supplies the start-up current to the primary-side controller, UCC3809. Operating current isDResistor R6 and capacitor C1 filter out leading-edge current spikes which are caused by the reverserecovery of the rectifier, equivalent capacitive loading on the secondary, and parasitic circuit inductances.C1 should be returned directly to the IC ground.DResistor R24 and capacitor C13 provide leading-edge blanking to the voltage spike resulting from theleakage inductance of the transformer. Failure to add these components would result in D4 and C5 peakrectifying to this spike voltage.DThe primary-side controller functions are supported by external circuitry such as resistors R3 and R1,combined with C9, which provide a charge and discharge path for the internal oscillator, setting the switchingfrequency for the converter.compensation.DTransistor Q4, resistors R20, and R22, along with dc voltage-blocking capacitor C15 provide slopeDCapacitor C3 programs the soft-start time and transistor Q2, biased from the resistor divider R2 and R4,is used to form an external shutdown.DCapacitors C2 and C4 are decoupling capacitors which should always be good quality low ESR/ESL typecapacitors placed as close to the IC pins as possible and returned directly to the IC ground reference.DThe gate-drive circuitry is composed of gate-drive resistor R5 (necessary for damping any oscillationsresulting from the input capacitance of Q3 and any parasitic stray inductance) and pull-down resistor R13(this resistor assures Q3 stays off in the event that U1 is removed from the circuit for any reason). ResistorR18 and diode D1 are used to accelerate the turnoff time of Q3, while still limiting the gate current duringswitch turnoff, thereby protecting the output stage of U1.DThe voltage-sense feedback loop is comprised of resistor-divider network R21, R17, and R16 (actually R21provides a series 50-Ω injection point for small-signal control-loop analysis). Feedback components R15and C16 provide the necessary gain and pole to stabilize the control loop, while R14 and R19 provide biascurrent to optocoupler Q1, and secondary-side error amplifier and voltage reference Z1. Capacitor C17helps to filter noise that may corrupt the exposed base terminal of the optocoupler while R9 provides theproper offset for the voltage-feedback signal to be summed with the current-sense signal and the slopecompensation at the FB pin of the UCC3809.3Theory of OperationWhen Q3 is turned on, T1 primary current increases linearly from zero. During this time, energy is stored in thegap of the transformer while the load current is supplied by the output-capacitor bank, C10 through C12, as wellas C14. When this primary current has increased to a value at which the voltage across R7, summed with thevoltage sense and slope compensation, exceeds the FB threshold voltage of one volt, Q3 is turned off. Thestored energy in T1 is now transferred to the secondary side, forward biasing D3, and also replenishing thecharge on the output-capacitor bank. 10-Watt Flyback Converter Using the UCC38095SLUU087BPeak current mode control responds immediately to line voltage changes and also provides over currentprotection to the switching device. A discontinuous current-mode controlled flyback converter does not have aright half plane zero and does not have the subharmonic-loop instability problems usually seen in the largeduty–ratio continuous–inductor current topologies. Although this converter does have peak-current limit, andsurvives a momentary short circuit on the output, an extended short circuit results in power supply failure.4UCC3809EVM–052 Operating InstructionsINPUTVOLTAGE DMM–+OUTPUTVOLTAGE DMM+–+INPUT POWERSUPPLY–J1J2J5DUTJ3+OUTPUTELECTRONICLOADJ4––+–+INPUTCURRENT DMMOUTPUTCURRENT DMMNOTES:1.Source power should be able to supply a minimum of 0.5 A at 75 V.NOTES:2Load should be able to sink up to 3 A with adequate power rating. Resistive loads with adequate ratings may be used.Figure 2. Connection Diagram for the UCC3809EVM–0521.Connect a dc power supply, capable of supplying up to 75 V at 0.5 A, to terminals J1 and J2. Notethat the positive terminal is connected to J1, the negative terminal is connected to J2. The powersupply should be set to its minimum-output voltage level and current limited to 0.5 A.2.Connect the load to J3 and J4. This load should be capable of handling 3 A at 3.3 V. The positiveterminal is connected to J3, the negative terminal is connected to J4.The output load should never exceed 3 A.Do not short the output terminals, power converter failure results.3.Turn on the electronic load.4.Turn on the input power supply. Starting with the input dc power supply at its minimum setting,slowly increase the voltage to the EVM operating range, 32 V to 75 V. The converter actuallybegins regulating when the dc supply is at approximately 24 V.The input voltage to the EVM should be no greater than 75 V at any time.610-Watt Flyback Converter Using the UCC3809SLUU087B5Determine the Maximum on Time (DMAX) for the Primary SwitchMaximum duty cycle occurs at minimum input voltage and maximum load. Programming DMAX clamps themaximum on–time of the switch. This maximum on–time should incorporate enough of a margin so that controlis well maintained at minimum input voltage. The UCC3809 has a programmable maximum duty-cycle clampup to 70%. Core saturation is prevented and discontinuous mode is ensured by establishing a dead time, usuallyapproximately 20% of the switching period. For this converter it is assumed DMAX is 40% and the clamp is setat 50% Setting a maximum duty cycle protects the magnetics from saturating during startup and brown outs.Selecting RT1 equal to RT2 results in 50% duty cycle, the value of CT is selected to satisfy the followingequation:1*27pF+141.919pFfSW 0.74 (RT1)RT2)where fSW is the desired switching frequency of the converter. A standard value of 150 pF results in a measuredswitching frequency of 380 kHz.6Determine the Turns Ratio of the Flyback Inductor By equating the volt–second on product, which is equal to the minimum input voltage, VIN(min), minus the drainto source forward voltage drop, VDS, multiplied by the maximum on–time, tON(max) = DMAXT, to the resetvolt-second product, equal to the output voltage, VOUT, added to the forward voltage drop of the output diode,VF, multiplied by the reset time, tRESET = (0.8–DMAX)T, the core is prevented from drifting up or down itshysteresis loop. The turns ratio of the transformer can be calculated by using this steady state volt–secondapproach:n+ǒVIN(min)*VDSǓ DMAX TǒVOUT)VFǓ ǒ0.8*DMAXǓ TThis application utilizes a turns ratio of 7, the primary consists of 14 turns while the secondary has 2 turns.7Primary InductanceMagnetic design is a major part of any switch-mode power supply. The flyback transformer is actually a coupledinductor, acting as an energy storage unit as well as performing the usual transformer functions. Crucialconsiderations include primary inductance, working flux density swing, gap length, winding scheme and wirediameter. The primary inductance, LP, for a discontinuous mode flyback converter can be calculated accordingto the following relationship:n LP+ƪǒVIN(min)*VDSǓ tON(max)ƫ2 T VOUT IOUT2where n is the assumed efficiency of the converter and IOUT is the output current. This converter design requiresa primary inductance of approximately 15 µH, typical. With a required output power of 10W and assuming 70%efficiency, the core should be sized to safely handle at least 14 W. The ferrite core should have high saturation,low residual flux density, and low losses. An RM5 core of 3F3 material proved to be suitable for this application.Hysteresis loss is minimized in this design by restricting the flux density to 800 gauss. Selecting a core materialwith high permeability is not crucial because the energy stored in the flyback transformer is actually stored inthe air gap. Gapping the core also reduces the residual flux density. The size of the air gap is calculated byapplying the following equation: 10-Watt Flyback Converter Using the UCC38097SLUU087BmO mR NP AE 10*2GAP+LPwhere µo is the permeability of free space equal to 4π×10–7 H/m and µR is the relative permeability of free space,which is air, equal to 1. AE is the effective core area measured in cm2 .28Soft StartSoft start is used to reduce transformer and output capacitor stress and to reduce the surge on the input circuitswhen the converter action starts. The considerable capacitance on the output lines should be charged slowlyso as not to reflect an excessive transient. Also, a wide initial pulse could result in saturation of the core, andvoltage overshoot on the output results if the inductor current is allowed to rise to a high value during start-up.To program this rise time, referred to as tSS, a 10-nF capacitor is chosen so that the 9.1-µA of maximuminternal-charging current from the SS pin of the UCC3809 delays complete start-up for approximately 2 ms (800cycles). When the voltage on this capacitor charges from 1 to 2 volts, the duty cycle is gradually increased fromnarrow pulses to normal operating conditions. If pulled below 0.5 V the output driver is inhibited and the ICreference is pulled low. The controller goes into its low start-up current of less than 100 µA. A simple NPNtransistor and two resistors allow the user to externally control this feature by application of a 5-V signal to J5as shown in Figure 1.9Self-Biasing WindingAt first power up, the UCC3809 has a very low start-up current of approximately 100-µA. This current can besupplied by a start-up resistor from the converter input line. But constant IC operating current, as high as 1.25mA at 400 kHz at 25_C, no load, in addition to the average gate-drive current from this source would beimpractical for efficiency reasons. Therefore, a bias winding is added to the Flyback inductor. This winding isessentially another secondary output but its return is common to the primary side. This winding must maintainvoltage above the undervoltage lockout (UVLO) threshold of the UCC3809 while supplying the IC with itsoperating current as well as the average gate-drive current, as mentioned above. The turns ratio for the biaswinding is determined using the same procedure as the main secondary, explained previously, except that VOUTis replaced with VDD. A small leading edge blanking RC circuit is placed at the output of the transformer to filterout the voltage spike due to leakage inductance of the transformer. A 100-V Schottky is used because thedevice’s working peak reverse voltage rating must be able to handle the reflected primary voltage added to thereflected main output voltage along with the VDD voltage that it is supplying. The bias winding capacitor mustbe able to maintain the IC input voltage above the UVLO threshold, VUVLO, while supplying both the operatingcurrent for the UCC3809, IVDD, and the average gate-drive current, IGATE(avg). A minimum value for CBOOT canbe calculated as:CBOOT+ǒIGATE(avg))IVDDǓ tSSǒVDD*VUVLOǓ10Selection of the Switching ElementThe MOSFET switch is selected to meet the drain to source voltage stress resulting from the maximum inputvoltage, VIN(max), the reflected secondary voltages, equal to the output voltage, VOUT, plus the output diodeforward voltage drop, VF, and the voltage spike due to the leakage inductance, assumed to be 30% of the inputvoltage:VDS(stress)+ƪǒVIN(max)Ǔ)N ǒVOUT)VFǓƫ)0.3 VIN(max)810-Watt Flyback Converter Using the UCC3809SLUU087BThe switch must be able to conduct the repetitive peak-primary current as determined by:IPEAK(primary)+ǒVIN(min)*VDSǓ tON(max)LPThe primary-current waveform of a discontinuous mode flyback converter is triangular in shape, therefore, itsroot mean square (rms) current is calculated by:IPRMS+IPEAK(primary)Ǹ3 ǸtON(max)TThe chosen device should also have a low RDS(on) value because the conduction losses of the device areproportional to the square of the primary rms current through the device. Selection of a device that has a peakcurrent rating of at least three times the peak-primary current usually ensures acceptably low conduction losses.PCONDUCTION+IPRMS RDS(on)Switching losses are the result of overlapping drain current and source voltage at turn off. The drain voltagebegins to rise only after the Miller capacitor of the device begins to discharge. This discharging time is a functionof the external gate resistance, RGATE, and the gate to drain Miller charge, QGD, as shown in the followingequation:2tMILLER+QGD RGATEVDD*VTHwhere VTH is the turn on threshold voltage of the gate.The power loss due to the external capacitance of the MOSFET also contributes to the total switching losses,which can be calculated as shown:ȡCOSS VDS(stress)2ȣPSWITCHING+fSW ȧ)VDS(stress) IPEAK(primary) tMILLERȧ2ȢDuring turn on there is no overlap of drain voltage and current because there is no current in a discontinuouscurrent-mode converter at turn on. Minimal losses also occur during the off-time of the FET due to the leakagecurrent:POFF(time)+ǒ1*DMAXǓ ILEAK VDS(stress)11Current SenseThe ground referenced sense resistor is selected such that the maximum peak-primary current trips the 1-V FBpin threshold when this current is 10% higher than its normal operating peak value at the minimum input voltage.This limits the peak-primary current in the event of an output short circuit. This resistor must have a power ratingto meet the IRMS2R requirement, where IRMS is the root mean square (rms) primary current. Because thisresistor defines the maximum peak-primary current, the input energy to the transformer is defined and equalto (1/2)LPIPEAK2 . This defined energy in a fixed frequency discontinuous-mode flyback results in a fixed outputpower. The advantage of current-mode control is that the output voltage is held constant despite changes in theinput voltage because the peak-primary current remains constant; the slope of this inductor current and its pulsewidth are adjusted. Leading edge spikes or noise are caused by the reverse recovery of the rectifier, equivalentcapacitive loading on the secondary, and parasitic circuit inductances. A small low pass RC filter is added tothe current-sense signal to filter out these spikes so the comparator does not assume an overload condition ispresent during switch turn on. To avoid excessive phase lag on the current-sense signal, the low pass filtercorner frequency is selected to be at least a decade above the switching frequency. 10-Watt Flyback Converter Using the UCC38099SLUU087B12Gate DriveAt 15 volts gate to source voltage the IRFR220 has a total gate charge of approximately 14 nC. The UCC3809is capable of sourcing 400-mA of peak-drive current which would result in a turn-on time of 35 ns. To limit thepeak current through the IC, an external resistor is placed between the totem-pole output of the IC and the gateof the MOSFET. The minimum value of this resistor is determined by:RGATE+VDD(min)*VSATIGATE(peak)This small-series resistor also damps any oscillations caused by the resonant tank of the parasitic inductancesin the traces of the board and the FET’s input capacitance. A pulldown resistor is added to the gate drive to insurethe MOSFET gate does not get charged to its turnon threshold during device start up. Adding a fast-switchingdiode and smaller value resistor in parallel with the gate resistor helps to control the current the IC needs to sinkduring turnoff and protects the output stage of the device. These components also help to reduce turnoff losseswhich tend to dominate the switching losses in discontinuous current-mode (DCM) converters.13Output DiodeThe output diode in a flyback converter is subject to large peak and rms current stresses. The 10-W flybackconverter described here has measured peak-secondary currents as high as 18 A with an rms value ofapproximately 7 A. Schottky diodes are recommended because of their low-forward voltage drop and the virtualabsence of minority carrier reverse recovery. The secondary-side Schottky rectifier was selected to meet theworking peak-reverse voltage, the peak repetitive-forward current, and the average forward current of theapplication. The working peak-reverse voltage, VREV, or blocking voltage, is calculated according to thefollowing equation:VREV+VIN(max))VRDS(on) 1)ǒVOUTǓNThe reflected peak-primary current constitutes the peak repetitive-forward current through the diode. Becauseall current to the output capacitor and load must flow through the diode, the average-forward diode current isequal to the steady-state load current. Power loss in the Schottky is the summation of the conduction lossesand reverse leakage losses. Conduction losses are calculated using the forward voltage drop across the diodeand the average-forward current. Reverse leakage losses are dependent upon the reverse-leakage current, theblocking voltage, and the on time of the FET.ǒǓ1010-Watt Flyback Converter Using the UCC3809SLUU087B14Output and Input CapacitorsOutput capacitors are selected based upon their capacitance value, equivalent series resistance (ESR),equivalent series inductance (ESL), and capacitor ripple current rating. The capacitance value controls the peakto peak output ripple voltage at the switching frequency. Assuming a linear decay of the capacitor voltage duringthe off time, during which the capacitor must supply the load current, the minimum value of the output capacitormay be calculated as follows:COUT+ǒT*tON(max)Ǔ IOUTVRIPPLEwhere VRIPPLE is the acceptable peak-to-peak output-voltage ripple. Unfortunately there are practical limitationsto how low a single stage output filter can reduce the ripple voltage and sometimes an extra LC filter stage isnecessary. This second-stage filter would also reduce the output high-frequency noise. Parasitic resistance andinductance in the output capacitors tends to make the ripple voltage much greater than expected based uponthe above equation. Using capacitors with the lowest possible ESR and ESL helps reduce the high-frequencyripple. The rms ripple current that the output capacitors experience is not the same as the secondary-side rmsoutput current; it is the ac portion of it. The secondary-side rms current is in the shape of a clipped sawtooth,or trapezoid, where as the output capacitor’s current waveform is in the shape of right triangle. Therefore, thetypical capacitor ripple current rating the output capacitors must meet is equal to:IRMS(cout)+IPEAK(sec) Ǹȱ4*3 ǒtRESETǓȳTtRESETȧȧǒTǓȧȧ12ȧȧȲȴwhere IPEAK(sec) is the peak-secondary current, and tRESET is equal to the off time of the switch. The sameselection criteria is used for the input capacitor, keeping in mind these capacitors must also be rated to handlethe maximum-input voltage.15Loop CompensationThe UCC3809 is a primary-side controller for use in isolated converters; therefore it does not contain an internalerror amplifier. The TLV431, a low-voltage adjustable-precision shunt regulator, is used on the secondary sidefor the feedback control loop. This regulator is ideal for low voltage supplies because the output voltage can beset to any value between its reference of 1.24 V and 6 V while operating from a lower voltage than the standardTL431.The output voltage is resistively divided and compared to the TLV431’s 1.24V reference voltage. When theresistively divided-converter output voltage rises above this threshold, the TLV431 drives the optocoupler diodeon which, in turn, drives the FB pin on the UCC3809 to its 1-V threshold, turning off the output driver. The reversehappens when the resistively divided-output voltage falls below the 1.24-V TLV431 reference voltage.The feedback loop needs to be closed around the error amplifier by adding a compensation network. Thenetwork components are selected so as to give the converter good dynamic response, acceptable line and loadregulation, and stability resulting from optimum closed-loop bandwidth. Before calculating the error-amplifiercompensation, the control to output gain, or transfer characteristic, along with the power-stage poles and zerosmust be determined. This is easily done using Lloyd Dixon’s Closing the Feedback Loop available in theUnitrode Power Supply Design Seminar SEM–700. The output capacitor, because of its parasitic ESR,contributes a zero in the frequency response at:FESR(zero)+12 p ESR COUT 10-Watt Flyback Converter Using the UCC380911SLUU087Bwhere COUT is the total capacitance of the output capacitor bank and ESR is the parallel combination of theoutput capacitor’s equivalent series resistance. The power stage contributes a pole located at:FPWRstg(pole)+2 p ROUT COUT2Note that this pole is load dependent. The control to output gain is calculated by the following equation:G+K IP+ǸROUT ǒLSEC fSWǓ22 ISC(sec)1*DMAXIK+PVCwhere ROUT is the load, LSEC is the primary inductance reflected to the secondary side, and ISC(sec) is thesecondary-side short-circuit current, VC is the reference voltage of the TLV431, or 1.24 V.Once the gain and corner frequencies have been determined, a bode plot can be constructed of thepre-compensated converter. Sampling theory limits the maximum-crossover frequency to one half the switchingfrequency. Practicality limits it even further; the system is unstable if the crossover frequency is more thanfSW/2πDMAX. For this design, the crossover frequency was set at approximately 1 kHz. The error amplifierrequires 38-dB of gain at this frequency. A pole is also needed to cancel the ESR zero. This pole is added adecade below the corner frequency of the ESR zero, resulting in an increase in low-frequency gain and adding45 degrees of phase margin. The error-amplifier compensation is determined using the following equation:EAGAIN+20 logǒǓRFRIwhere RI is equal to R21 + R17, referring to Figure 1, the feedback resistor, RF, shown as R15 in Figure 1, iseasily determined. The feedback capacitor (shown as C16 on the schematic), which, when combined with thefeedback resistor adds the ESR canceling pole, is calculated using:CF+12 p FP RFwhere FP is set at one-tenth FESR (zero).The resultant phase margin is approximately 100 degrees.1210-Watt Flyback Converter Using the UCC3809SLUU087B16Slope CompensationAll peak current-mode converters benefit from slope compensation, which is added to the current-sense signal,to cancel the error introduced by sensing peak current instead of average current. The amount of inductor downslope added to the FB pin was designed to be approximately 0.5 and the emitter follower configuration was used.The step-by-step calculations are beyond the scope of this design note. Application Note U–111 (TI LiteratureNumber SLUA111) is an excellent reference for slope compensation design techniques.17Silkscreen+48 INR11J1R10+OUTC10C11J3C12C6D2J248 INC18C7D3R7Q3R24C13C14D4C5J4OUTJ5SHUTDOWNNOTE:High–temperature component. See EVM Warnings and Restrictions at the back of this document.18ConclusionThis design procedure addresses the major considerations and calculations used in the design of a 10-Wdiscontinuous-mode flyback converter. These include maximum duty cycle, inductor turns ratio, core selection,primary inductance, gate drive, and current sensing. Major component selections such as switching element,output diode, and output capacitors were also discussed as well as control loop compensation. This converteris available as an evaluation module from Texas Instruments.19ReferencesBill Andreycak, Practical Considerations in High Performance MOSFET, IGBT and MCT Gate Drive Circuits,Unitrode Application Note U–137, (TI Literature Number SLUA156), Unitrode Applications Handbook IC#1051,1997.Bill Andreycak, Practical Considerations in Current Mode Power Supplies, Unitrode Application Note U–111, (TILiterature Number SLUA111), Unitrode Applications Handbook IC#1051, 1997.Keith Billings, Switchmode Power Supply Handbook, McGraw–Hill, Inc.,19.Lloyd Dixon, Jr., Closing the Feedback Loop, Unitrode Power Supply Design Seminar Manual SEM–1100, 1996. 10-Watt Flyback Converter Using the UCC380913SLUU087BDYNAMIC WARNINGS AND RESTRICTIONSIt is important to operate this EVM within the input voltage range of 32.0 V to 75.0 V, and the output voltage rangeof 3.218 V to 3.383 V, and the load current range of 0.0 A to 3.0 A.Exceeding the specified input range may cause unexpected operation and/or irreversible damage to the EVM.If there are questions concerning the input range, please contact a TI field representative prior to connectingthe input power.Applying loads outside of the specified output range may result in unintended operation and/or possiblepermanent damage to the EVM. Please consult the EVM User’s Guide prior to connecting any load to the EVMoutput. If there is uncertainty as to the load specification, please contact a TI field representative.During normal operation, some circuit components may have case temperatures greater than 50°C. The EVMis designed to operate properly with certain components above 50°C as long as the input and output ranges aremaintained. These components include but are not limited to linear regulators, switching transistors, passtransistors, and current sense resistors. These types of devices can be identified using the EVM schematiclocated in the EVM User’s Guide. When placing measurement probes near these devices during operation,please be aware that these devices may be very warm to the touch.Mailing Address:Texas InstrumentsPost Office Box 655303Dallas, Texas 75265Copyright 2001, Texas Instruments Incorporated1410-Watt Flyback Converter Using the UCC3809IMPORTANT NOTICE
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